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MC33065DW-H View Datasheet(PDF) - ON Semiconductor

Part NameDescriptionManufacturer
MC33065DW-H HIGH PERFORMANCE DUAL CHANNEL CURRENT MODE CONTROLLERS ON-Semiconductor
ON Semiconductor ON-Semiconductor
MC33065DW-H Datasheet PDF : 18 Pages
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MC34065H, L
OPERATING DESCRIPTION
The MC34065H,L series are high performance, fixed
frequency, dual channel current mode controllers
specifically designed for OffLine and dctodc converter
applications. These devices offer the designer a cost
effective solution with minimal external components where
independent regulation of two power converters is required.
The Representative Block Diagram is shown in Figure 15.
Each channel contains a high gain error amplifier, current
sensing comparator, pulse width modulator latch, and totem
pole output driver. The oscillator, reference regulator, and
undervoltage lockout circuits are common to both
channels.
Oscillator
The unique oscillator configuration employed features
precise frequency and duty cycle control. The frequency is
programmed by the values selected for the timing
components RT and CT. Capacitor CT is charged and
discharged by an equal magnitude internal current source
and sink, generating a symmetrical 50 percent duty cycle
waveform at Pin 2. The oscillator peak and valley thresholds
are 3.5 V and 1.6 V respectively. The source/sink current
magnitude is controlled by resistor RT. For proper operation
over temperature it must be in the range of 4.0 kΩ to 16 kΩ
as shown in Figure 1.
As CT charges and discharges, an internal blanking pulse
is generated that alternately drives the center inputs of the
upper and lower NOR gates high. This, in conjunction with
a precise amount of delay time introduced into each channel,
produces well defined nonoverlapping output duty cycles.
Output 2 is enabled while CT is charging, and Output 1 is
enabled during the discharge. Figure 2 shows the Maximum
Output Duty Cycle versus Oscillator Frequency. Note that
even at 500 kHz, each output is capable of approximately
44% ontime, making this controller suitable for high
frequency power conversion applications.
In many noise sensitive applications it may be desirable to
frequencylock the converter to an external system clock.
This can be accomplished by applying a clock signal as
shown in Figure 17. For reliable locking, the freerunning
oscillator frequency should be set about 10% less than the
clock frequency. Referring to the timing diagram shown in
Figure 16, the rising edge of the clock signal applied to the
Sync input, terminates charging of CT and Drive Output 2
conduction. By tailoring the clock waveform symmetry,
accurate duty cycle clamping of either output can be
achieved. A circuit method for this, and multiunit
synchronization, is shown in Figure 18.
Error Amplifier
Each channel contains a fullycompensated Error
Amplifier with access to the inverting input and output. The
amplifier features a typical dc voltage gain of 100 dB, and
a unity gain bandwidth of 1.0 MHz with 71° of phase margin
(Figure 5). The noninverting input is internally biased at 2.5
V and is not pinned out. The converter output voltage is
typically divided down and monitored by the inverting input
through a resistor divider. The maximum input bias current
is 1.0 μA which will cause an output voltage error that is
equal to the product of the input bias current and the
equivalent input divider source resistance.
The Error Amp output (Pin 5, 12) is provided for external
loop compensation. The output voltage is offset by two
diode drops (1.4 V) and divided by three before it connects
to the inverting input of the Current Sense Comparator. This
guarantees that no pulses appear at the Drive Output (Pin 7,
10) when the error amplifier output is at its lowest state
(VOL). This occurs when the power supply is operating and
the load is removed, or at the beginning of a softstart
interval (Figures 20, 21).
The minimum allowable Error Amp feedback resistance
is limited by the amplifier’s source current (0.5 mA) and the
output voltage (VOH) required to reach the comparator’s
1.0 V clamp level with the inverting input at ground. This
condition happens during initial system startup or when the
sensed output is shorted:
Rf(min)
3.0 (1.0 V) ) 1.4 V = 8800
0.5 mA
Ω
Current Sense Comparator and PWM Latch
The MC34065 operates as a current mode controller,
whereby output switch conduction is initiated by the
oscillator and terminated when the peak inductor current
reaches the threshold level established by the Error
Amplifier output. Thus the error signal controls the peak
inductor current on a cyclebycycle basis. The Current
Sense ComparatorPWM Latch configuration used ensures
that only a single pulse appears at the Drive Output during
any given oscillator cycle. The inductor current is converted
to a voltage by inserting a groundreferenced sense resistor
RS in series with the source of output switch Q1. This
voltage is monitored by the Current Sense Input (Pin 6, 11)
and compared to a level derived from the Error Amp output.
The peak inductor current under normal operating
conditions is controlled by the voltage at Pin 5, 12 where:
Ipk =
V(Pin 5, 12) 1.4 V
3 RS
Abnormal operating conditions occur when the power
supply output is overloaded or if output voltage sensing is
lost. Under these conditions, the Current Sense Comparator
threshold will be internally clamped to 1.0 V. Therefore the
maximum peak switch current is:
Ipk(max) =
1.0
RVS
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